1. Technical Field
The present disclosure relates to a control device for a rectifier of a switching converter.
2. Description of the Related Art
Resonant converters are a large class of forced switching converters characterized by the presence of a half-bridge or a full-bridge circuit topology. In the half-bridge version, for example, the switching elements comprise a high-side transistor and a low-side transistor connected in series between an input voltage and ground. A square wave having a high value corresponding to the power supply voltage and a low value corresponding to ground may be generated by conveniently switching the two transistors.
The square wave generated by the half-bridge is applied to the primary winding of a transformer by means of a resonant circuit which comprises at least one capacitor and one inductor. The secondary winding of the transformer is connected with a rectifier circuit and a filter to provide an output direct voltage depending on the frequency of the square wave.
At present, one of the resonant converters most widely used is the LLC resonant converter. This name derives from the fact that the resonant circuit employs two inductors (L) and a capacitor (C). A schematic circuit of an LLC resonant converter is shown in FIG. 1 and comprises a half-bridge of MOSFET transistors M1 and M2, with respective body diodes Db1 and Db2, coupled between an input voltage Vin and ground GND and driven by a driver circuit 3. The common terminal between transistors M1 and M2 is connected to a resonant network 2 comprising a series of a first inductance Lr, a second inductance Lm and a capacitor Cr; the inductance Lm is connected in parallel to a transformer 10 comprising a secondary winding connected to the parallel of a capacitor Co and a resistance Rout by means of the rectifier diodes D1 and D2. The output voltage Vo of the resonant converter is the voltage across said parallel, while the output current Io flows through the resistance Rout.
These resonant converters are characterized by a high conversion efficiency (>95% is easily achievable), an ability to work at high frequencies, low generation of EMI (Electro-Magnetic Interference).
In current types of converter circuits, a high conversion efficiency and high power density are desired, as in the case, for example, of the AC-DC adaptors of notebooks. LLC resonant converters are at present the converters that best meet such requirements.
However, the maximum efficiency achievable is limited by the losses in the rectifiers on the secondary side of the converter, which account for over 60% of total losses.
It is known that in order to significantly reduce the losses connected to secondary rectification, recourse can be made to the so-called “synchronous rectification” technique, in which rectifier diodes are replaced by power MOSFETs, with a suitably low on-resistance, such that the voltage drop across it is significantly lower than that across the diode; and they are driven in such a manner as to be functionally equivalent to the diode. This technique is widely adopted in traditional converters, especially in flyback and forward converters, for which there also exist commercially available dedicated integrated control circuits. There is an increasingly pressing desire to adopt this technique in resonant converters as well, in particular in LLC converters, in order to enhance their efficiency as much as possible.
FIG. 2 shows the converter of FIG. 1 in the version with secondary synchronous rectifiers; in this case, in the place of diodes D1 and D2 there are two transistors SR1 and SR2, suitably driven by two signals G1S and G2S deriving from a driver 80, and connected between the terminals of the two parts of the center-tapped secondary winding connected to ground GND, while the parallel of Co and Rout is disposed between the center tap of the secondary winding and ground GND. From a functional viewpoint there is no difference, as compared to the schematic in FIG. 1.
The transistors SR1 and SR2 have respective body diodes Dbr1 and Dbr2, and are both driven by a synchronous rectifier driver 80. The output voltage Vo of the resonant converter is the voltage across said parallel, while the output current Io flows through the resistance Rout.
In operation, the transistors SR1 and SR2 are driven in such a manner to be alternatively turned-on at a certain frequency by the synchronous rectifier driver. When the body diode Dbr1, Dbr2 of one of the transistors SR1, SR2 starts conducting the relative transistor is turned-on, while when the current is approaching zero the transistor is turned-off; in this way the use of the transistors SR1, SR2 causes a lower voltage drop than the use of the diodes D1, D2 and the power dissipation is reduced.
Particularly, as is shown in FIG. 3, a phase A is activated when the voltage Vdvs between the drain and source terminals of one of the transistor SR1, SR2, for example the voltage Vdvs1 of the transistor SR1, is lower than a voltage value of 0.7V the relative body diode Dbr1 starts conducting; then when the voltage Vdvs falls under a turn-on threshold voltage VTH—ON and after a fixed delay time period TPD—ON, when the voltage Vdvs is maintained under the turn-on threshold voltage VTH—ON, the transistor SR1 is turned on from the driver.
After the turn on of the transistor SR1, in a phase B, the voltage Vdvs1 has a value of Vdvs=−Rdson×Isr, wherein Rdson is the on resistance of the transistor SR1 and Isr is the current flowing through the electric path between the center-tap CT of the secondary winding of the transformer and ground GND.
When the voltage Vdvs has a value higher than a second threshold voltage VTH—OFF, the transistor SR1 is turned off by the rectifier driver 80. The respective body diode Dbr1 conducts again and the voltage Vdvs goes negative; when the voltage Vdvs1 reaches the value of 1.4V, the drive circuit relative to the transistor SR2 is enabled.
However, the voltage Vdvs1, Vdvs2 depends on the parasitic elements of the source and drain terminal of the transistor SR1, SR2 and of the path of printed circuit board (PCB) from the drain terminal of the transistor SR1, SR2 and the terminal of the secondary winding. Particularly, the voltage Vdvs1, Vdvs2 depends on the parasitic inductances Lsource and Ldrain associated to the source and drain terminal of the transistor SR1, SR2 and on the parasitic inductance Ltrace relative to the path of printed circuit board (PCB) from drain terminal of the transistor SR1 or SR2 and the terminal of the secondary winding, therefore
      Vdvs    =                            -          Rdson                ×        Isr            -                        (                      Ldrain            +            Lsource            +            Ltrace                    )                ×                              ∂            Isr                                ∂            t                                ,where when Vdvs and Isr are indicated as one of the voltages Vdvs1, Vdvs2 and the currents Isr1, Isr2; the parasitic inductances make the sensed voltage Vdvs1, Vdvs2 different from the ideal voltage drop value on Rdson.
The presence of the parasitic inductances Ldrain, Lsource and Ltrace determines an undesired earlier turn-off of the transistors SR1, SR2 as shown in FIG. 4 where the drain-source voltage Vdvs and the desired voltage Vdvs-ideal are shown. The residual conduction time Tdiode of the body diode Dbr1 or Dbr2 increases, causing a loss of efficiency (indicate with LE in FIG. 4) due to the higher voltage drop across the body diode Dbr1 or Dbr2.
For example, a typical starting body diode residual conduction time Tdiode could be of 1 micro second, while a typically desired time Tdiode value is 60 nanoseconds.
A known technique to avoid the earlier turn-off of the transistors SR1, SR2 (FIG. 5) is to compensate the time anticipation due to the parasitic inductances by adding an RC filter downstream the rectifier driver 80 and before the transformer 10. The RC filter comprises an external capacitor Ccomp and a tunable resistor Rd. Current inversion has to be avoided to prevent converter malfunctions and failure.
This solution has the advantages of providing a simple architecture with a consequent low cost in term of silicon area and good performance.
However, external components to optimize efficiency are needed. Furthermore the RC compensation of the of the parasitic inductances Ldrain, Lsource and Ltrace may cause a delay to turn on the transistors SR1, SR2; a bypass diode arranged in parallel to the resistor Rd eliminates this turn-on time delay. Furthermore, a resistor, of the value of about 100-200Ω, arranged in series to the bypass diode (not shown in FIG. 4) is typically used to limit the current Isr1,2 in the case wherein the voltage Vdvs goes excessively under ground GND.
Also the efficiency of the solution of prior art is dependent on the residual conduction time of the body diodes Dbr1, Dbr2 which in turn depends on the on resistances of the transistors SR1, SR2, on the parasitic elements of the transistors and the printed circuit board wherein the transistors are implemented, on the temperature and on the slew rate of the current flowing through the transistor.
Furthermore, during fast load transient, for example when the LLC converter works above resonance, there is the risk of an inversion of the current flowing through the transistor SR1, SR2 and consequently of a malfunction and even failure of the converter.